The invention relates to a filter assembly of a front-end module used in mobile communication systems, and more specifically to a filter assembly capable of balanced to unbalanced conversion.
Currently, dual-band, tri-band and quad-band mobile phones are available for operating in different mobile communication systems. Dual-band mobile phones switch between the operating frequencies of 900 MHz and 1800 MHz, tri-band mobile phones switch between the operating frequencies of 900 MHz, 1800 MHz and 1900 MHz, and quad-band mobile phones switch between the operating frequencies of 900 MHz, 1800 MHz, 1900 MHz and 850 MHz. The quad-band communication system is characterized by support of media applications, high speed connectivity and fast audio and image downloads. Additionally, quad-band mobile phones are compatible with all mobile communication systems. Consequently, it is desirable to have a front-end module for accommodating to the quad-band mobile communication systems.
FIG. 1 shows a conventional front-end module in a mobile communication system. As shown in FIG. 1, the front-end module 10 comprises a duplexer 12, a power amplifier 22, a low noise amplifier 24 and an antenna 20. The duplexer 12 comprises two bandpass filters 14 and 16, a 90° phase shifter 18, an output terminal 13 coupled between the bandpass filter 14 and the power amplifier 22, an input terminal 15 coupled between the bandpass filter 16 and the low noise amplifier 24, and an antenna terminal 17 connected to the antenna 20.
A transmitted signal is amplified by the power amplifier 22 to the duplexer 12 via the output terminal 13. The bandpass filter 14 proceeds to allow the signal within a certain frequency band to pass therethrough and the antenna 20 then transmits the passed signal via the antenna terminal 17. Similarly, when receiving a signal, the antenna 20 feeds the received signal to the 90° phase shifter 18 via the antenna terminal 17 and the signal is then outputted to the bandpass filter 16, which applies the received signal to the low noise amplifier 24 via the input terminal 15. The low noise amplifier 24 then filters the noise in the low frequency (LF) signal passing through the passband filter 16 and amplifies the signal. In order to eliminate the quality degrading disturbance in the signal passing through the bandpass filter 16, which stems from the signal passing through the bandpass filter 14, a 90° phase shifter 18 is typically disposed therebetween to separate the transmitted and received signals by the difference in signal phases. The signal, however, must be transformed from an unbalanced signal to a balanced signal when outputted to the antenna terminal 17 via the output terminal 13 or inputted from the antenna 17 to the input terminal 15. Hence, an unbalanced to balanced conversion transformer (Balun) is required to be disposed before the bandpass filter 14 or after the bandpass filter 16 in the signal transmission path to reject the noise.
In recent years, when making filters and duplexers used in RF communication systems, the piezoelectric thin film process is typically employed in manufacturing ultrasonic components. The conventional piezoelectric thin film acoustic component can be roughly classified as a film bulk acoustic resonator (FBAR) or a solidly mounted resonator (SMR) depending on structure. SMR is supported by a Bragg reflector and FBAR is manufactured in Microelectromechanical (MEMS) surface micromachining or bulk micromachining to empty the parts below the lower electrode or supporting layer to enable the thin film structure to conform to the total reflection boundary condition of acoustic waves.
FIG. 2A is a schematic diagram of a film bulk acoustic resonator (FBAR) in a stacked crystal filter (SCF) arrangement. As depicted in FIG. 2A, the FBAR 30 comprises an input electrode 32, an output electrode 34, a grounded electrode 36, an upper piezoelectric layer 31, and a lower piezoelectric layer 33. The input electrode 32 receives a signal from the input terminal 35 and the upper piezoelectric layer 31 then generates a bulk acoustic wave to the lower piezoelectric layer 33 in response to the signal excitation, thereby a resonance is generated between the input electrode 32 and the output electrode 34. The output electrode 34 then outputs the signal to an output terminal 37. The FBAR 30 is only used for unbalanced signal transmission since the input and output signals share the grounded electrode 36.
In order to compensate for the defect of the FBAR 30, another FBAR in a SCF arrangement is provided as shown in FIG. 2B. The FBAR 40 comprises an input electrode 42, two output electrodes 46 and 48, a grounded electrode 44, an upper piezoelectric layer 41, a lower piezoelectric layer 43 and a dielectric layer 50. While the transmission principle is the same as the described operation, the FBAR 40, however, is capable of both balanced and unbalanced signal transmission due to the insulation of the dielectric layer 50. Thus, the unbalanced signal only exists at the input terminal 45 of the input electrode 42 and the grounded terminal of the grounded electrode 44 whereas the balanced signal is outputted to the output terminals 47 and 49 by the output electrodes 46 and 48.
FIG. 2C shows a conventional FBAR in a coupled resonator filter (CFR) arrangement. The FBAR 60 of FIG. 2C comprises an input electrode 62, two output electrodes 66 and 68, a grounded electrode 64, an upper piezoelectric layer 61, a lower piezoelectric layer 63, a plurality of first coupling layers 72 and a plurality of second coupling layers 74. The operating principle thereof is the same as the previously described operation because the FBAR 60 is similar to the FBAR 40 except that the dielectric layer 5Q is replaced with the interleaved first coupling layer 72 and the second coupling layer 74. The first coupling layer 72 and the second coupling layer 74 comprise different material having different acoustic impedance but the same thickness of a quarter-wavelength. Thus, the unbalanced signal only exists at the input terminal 65 of the input electrode 62 and the grounded terminal of the grounded electrode 64 whereas the balanced signal is outputted to the output terminals 67 and 69 by the output electrodes 66 and 68.
FIG. 3A shows a schematic diagram of the frequency response of the FBAR 40 in FIG. 2B. As shown in FIG. 3A, there are three resonant modes in the frequency response of FBAR 40. Generally speaking, there is only a passband in a common bandpass filter and the operating band of the quad-band mobile communication system is from 850 MHz to 1900 MHz, thus the modes outside the 1400 MHz to 1800 MHz frequency range cannot reject noise effectively, resulting in signal quality degradation. Hence, the FBAR 40 and FBAR 60 cannot meet the low noise requirement of the mobile communication systems.
FIG. 2D shows a solidly mounted resonator (SMR) in SCF arrangement and FIG. 3B is the frequency response thereof. As shown in FIG. 2D, the SMR 80 comprises an input electrode 82, two output electrodes 86 and 88, a grounded electrode 84, an upper piezoelectric layer 81, a lower piezoelectric layer 83, a dielectric layer 90, a plurality of first reflective layers 92, a plurality of second reflective layers 94 and a substrate 96. The operating principle of the SMR 80 is substantially similar to the FBAR 40 except that a plurality of first and second reflective layers 92 and 94 are disposed under the lower surface of the output electrode 88 for support, wherein the interleaved first reflective layer 92 and the second reflective layer 94 formed on the substrate 96 are acoustic reflectors and made of different materials having different acoustic impedance. The thickness of the first reflective layer 92 and the second reflective layer 94 is a quarter acoustic wavelength and thus when the acoustic wave progresses to the first and second reflective layers 92 and 94, a Bragg reflection which approximates a total reflection is formed and the resonant energy is then maintained in the SMR 80. The unbalanced signal only exists at the input terminal 85 of the input electrode 82 and the grounded terminal of the grounded electrode 84 whereas the balanced signal is outputted to the output terminals 87 and 99 by the output electrodes 86 and 88. FIG. 3B shows a frequency response curve 140 of the SMR 80 when the first reflective layer 92 and the second reflective layer 94 are made of tungsten (W) and SiO2 respectively, and a frequency response curve 130 when the first reflective layer 92 and the second reflective layer 94 are made of AlN and SiO2 respectively. Because the acoustic impedance ratio between AlN and SiO2 is smaller than that between W and SiO2, a narrower reflection bandwidth having better noise rejection performance in the bandpass filter is obtained. The reflectivity of the reflective layers with smaller acoustic impedance ratio, however, is worse, thus there exists a need for more layers. Consequently, the propagation route of the acoustic wave is increased and the Q value thereof is also degraded due to the transmission loss.